1. Field of the Invention
The present invention relates to a low noise amplifier, and, particularly, to a low noise amplifier with a wide dynamic range, which is used as a high-frequency amplifier for a radio communication device, an input amplifier for an A/D converter, or the like.
2. Description of the Related Art
As a conventional low noise amplifier, there is a cascode-connected amplifier disclosed in Unexamined Japanese Patent Application KOKAI Publication No. 2003-289226. The cascode-connected amplifier is known as a preferable circuit for adaptation to a wide-band amplifier, which is hardly affected by the parasitic capacitor of an input transistor.
“HF Low Noise Amplifiers with Integrated Transformer Feedback” (ISCAS 2002, vol. 2, pp. II-815-II-818, May 2002 written by K. van Hartingsveldt, M. H. L. Kouwenhoven, C. J. M. Verhoeven, A. N. Burghartz) discloses a low noise amplifier having a double negative feedback network, which includes a transformer and a resistor. The low noise amplifier having the double negative feedback network is an excellent circuit, which can achieve a low noise figure, a stable gain and satisfactory input impedance matching.
The contents of Unexamined Japanese Patent Application KOKAI Publication No. 2003-289226 and the contents of “HF Low Noise Amplifiers with Integrated Transformer Feedback” are incorporated therein by reference.
It is theoretically possible to realize a low noise amplifier that has a wide dynamic range and operates with low power consumption by providing a transformer feedback cascode type low noise amplifier (Transformer Feedback Cascode LNA: TFC-LNA), which includes a combination of the cascode-connected LNA having the double negative feedback network, with a high feedback loop gain. However, the trade-off between providing a high feedback loop gain and increasing the cut-off frequency of the feedback loop gain raises a problem such that if a high feedback loop gain is held up to a high frequency band, applying the conventional compensation approach to the LNA cannot provide sufficient phase compensation, resulting in easy oscillation, so that the LNA does not serve as an amplifier.
A TFC-LNA as shown in FIG. 20, which employs the dominant-pole compensation method, generally used to suppress oscillation, is one example of the conventional LNA. This LNA includes transistors having a gain bandwidth (transient frequency) fT of 8 GHz, and operates on a DC supply voltage VDC of 10 V supplied from a DC voltage source DCS.
A signal source 1 with an output impedance R of 50Ω is connected to an input terminal IN via a DC cut-off capacitor 2. The input terminal IN is connected with the hot side of the primary winding of the transformer 3, which has a turn ratio of, for example, 1:2.
The cold side of the primary winding of the transformer 3 is connected to the base of an NPN type transistor 4, which is the input node NI of the cascode amplifier. The base of the transistor 4 is further connected with the positive electrode of a biasing DC voltage source 5 via a choke coil 6.
The collector of the transistor 4 is connected with the emitter of an NPN type transistor 7. The base of the transistor 7 is connected to the positive electrode of a biasing DC voltage source 8, and is AC-grounded. The transistors 4 and 7 are cascode-connected, and the collector of the transistor 7 is connected with one end of a resistor 9, which becomes a load of the cascode amplifier. The other end of resistor 9 is supplied with the DC supply voltage VDC from the DC voltage source DCS.
A node between the resistor 9 and the collector of the transistor 7 is an output node NO from which the output signal of the cascode amplifier is output, and is connected to the base of an NPN type transistor 10, i.e., the input side of an emitter follower. The collector of the transistor 10 is supplied with the DC supply voltage VDC from the DC voltage source DCS. The transistor 10, together with a constant current source 18, constitutes the emitter follower, and operates as the output buffer of the LNA.
A capacitor 11 for phase compensation is connected between the base of the transistor 10, i.e., an output node NO of the cascode amplifier, and the positive electrode of the DC voltage source DCS, i.e., the AC ground. The resistor 9 and the capacitor 11 provide a dominant pole for the feedback loop gain, and work for low-pass filtering of the output signal of the cascode amplifier. The emitter of the transistor 10, i.e., an output terminal OUT of the LNA, is connected with a load 13 of the LNA via a capacitor 12. The load 13 comprises a resistor of, for example, 5 kΩ.
The cold side of the secondary winding of the transformer 3 is connected to the emitter of the transistor 10 or the output terminal OUT of the LNA via a capacitor 14. The hot side of the secondary winding of the transformer 3 is connected to the ground. An output voltage signal applied to the secondary winding of the transformer 3 is transferred to the primary winding side of the transformer 3 by electromagnetic coupling, and is fed back in series to the input voltage signal, thereby forming a first negative feedback network of the LNA.
A resistor 16 and a DC cut-off capacitor 17 are connected between the emitter of the transistor 10 or the output terminal OUT of the LNA and the hot side of the primary winding of the transformer 3 or the input terminal IN of the LNA for shunt feedback of the output signal. This resistor 16 constitutes a second negative feedback network of the LNA. The emitter of the transistor 10 is connected with a constant current source 18 to provide the operating current of the emitter follower.
In this example, the operating current of the emitter follower is set to about 12 mA and a gain of 200 (46 dB) or so is set for the voltage gain of the cascode amplifier, so that the maximum feedback loop gain of the LNA takes a high value equal to or greater than 40 dB.
The voltage gain of the LNA is theoretically given by a turn ratio 1:N of the transformer 3. In this example, the turn ratio of the transformer 3 is 1:2, so that the voltage gain of the LNA is about 6 dB. The transformer 3 is of a commercially available type, which has a loss of about 1.0 dB and a passband of about 3 to 200 MHz. The optimal resistance value of the resistor 16 to be the feedback resistor is theoretically given by an equation (N+1)R where R is the input impedance determined by the specification of the LNA and N is the turn ratio of the transformer 3. In this example, a typical value, 50Ω, is set as the input impedance R, and the resistance value of the resistor 16 is set to 150Ω.
As described above, the resistor 9 and the capacitor 11 work to produce a dominant pole in the transfer function of the feedback loop gain, thereby achieving phase compensation of the LNA. When phase compensation is carried out to provide the feedback loop gain observed at the base of the transistor 4 with a phase margin of about 45 degrees, the required capacitance of the capacitor 11 is equal to or greater than 140 pF. From the viewpoint of the cost restriction, it is not possible to form a capacitor with such a large capacitance on an integrated network. This requires that the capacitor 11 should be an external part, which disadvantageously leads to an increase in the number of parts and an increase in the printed circuit board area. This is one drawback of the dominant-pole compensation method.
FIG. 21 is a diagram showing the feedback loop gain of the LNA shown in FIG. 20, and has the results of simulated measurement of the feedback loop gain observed at the base of the transistor 4, plotted on a Bode diagram.
The feedback loop gain of the LNA, as shown in FIG. 21, takes a maximum value of about 44 dB at a frequency of 360 kHz or so, and drops down to 0 dB at about 190 MHz. The phase margin is 45 degrees, and the gain margin is about 5 dB. The −3 dB cut-off frequency at which the feedback loop gain starts to attenuate is about 1.1 MHz. It is apparent that the LNA keeps its high dynamic range intact only within a frequency band under a few MHz.
According to the dominant-pole compensation method, phase compensation is carried out in such a way that the feedback loop gain decreases at −20 dB/dec. The upper frequency limit of the passband at which the transformer 3 in FIG. 20 shows substantially ideal behavior is approximately 200 MHz, and in a high frequency band equal to or higher than the frequency, deterioration of the phase margin originated from the parasitic capacitor or the like of the transformer 3 becomes prominent in addition to the dominant-pole originated attenuation of the phase margin.
When the maximum feedback loop gain is set to a high value equal to or greater than 40 dB, therefore, the cut-off frequency should be set to 200 MHz/2 dec (=100) or 2 MHz or lower in taking this approach to compensate with a satisfactory phase margin. The band where a high feedback loop gain can be maintained is limited to a frequency significantly lower than the upper limit of the transformer passband, so that the LNA, if used for high frequency applications, cannot demonstrate a substantial performance. This is another drawback of the dominant-pole compensation method.
FIG. 22 shows simulation results which represent a third order intercept point (hereinafter referred to as “IIP3”) of the LNA shown in FIG. 20, the abscissa representing the frequency (MHz) while the ordinate represents IIP3 (dBm). In the simulation of the IIP3 characteristic of the LNA, two tone signals with power of −50 dBm at a frequency apart by 10 kHz from the measuring frequency are used as inputs.
As shown in FIG. 22, it is apparent that at 10 MHz, the IIP3 characteristic is deteriorated by 20 dB or greater from the maximum value of 42 dBm. The deterioration of the IIP3 characteristic has occurred in accordance with the attenuation of the feedback loop gain of the LNA. In general, in a negative feedback amplifier, as the feedback loop gain drops, the IIP3 value decreases too. In the LNA using the dominant-pole compensation method, as mentioned above, it is difficult to keep a high feedback loop gain at a high frequency band. Therefore, the LNA in FIG. 20 is not suitable for use as a high-frequency LNA, which demands a wide dynamic range.
The LNA using a Miller compensation method may be used instead of the dominant-pole compensation method shown in FIG. 20. In the dominant-pole compensation method, the capacitor 11 is connected to the output node NO of the cascode amplifier or between the base and collector of the transistor 10 for phase compensation, whereas in the Miller compensation method, a capacitor for phase compensation is connected between the output node NO of the cascode amplifier and the input node NI of the cascode amplifier or the base of the transistor 4.
Given that the capacitance of the phase compensation capacitor is C and the voltage amplification of the cascode amplifier is β, the phase compensation capacitor connected by the Miller compensation method works to present substantially the same effect as that in a case where a shunt capacitor having the capacitance (β-1)C is connected to the input node NI of the cascode amplifier. Therefore, the Miller compensation method can generally carry out phase compensation using a capacitor with a small capacitance as compared with the dominant-pole compensation method.
In the Miller compensation method, however, adding the phase compensation capacitor forms a new signal path from the input node NI to the output node NO, which passes through the phase compensation capacitor. Accordingly, a zero appears in the feedback loop transfer function at a high frequency band, and attenuation of the feedback loop gain becomes moderate in the vicinity of the zero. This shifts the cross-over frequency of the feedback loop gain to the high frequency side, resulting in reduction in phase margin.
In case of carrying out phase compensation by making the capacitance of the phase compensation capacitor larger to reduce the cut-off frequency of the feedback loop gain, the zero of the feedback loop transfer function shifts to the low frequency side at the same time, so that the cross-over frequency falls insufficiently, thus disabling the stabilization of the LNA. While it is possible to prevent the cross-over frequency from becoming higher by reducing the feedback loop gain, the amount of feedback is reduced, thus deteriorating the IIP3 characteristic. The above situations make it difficult to achieve phase compensation of the LNA having a high dynamic range.